Texas Instruments LM3429 User manual

LM3429

LM3429 LM3429Q1 N-Channel Controller for Constant Current LED Drivers

Literature Number: SNVS616F

LM3429

LM3429Q1

January 19, 2010

N-Channel Controller for Constant Current LED Drivers

General Description

The LM3429 is a versatile high voltage N-channel MosFET

controller for LED drivers . It can be easily configured in buck,

boost, buck-boost and SEPIC topologies. This flexibility,

along with an input voltage rating of 75V, makes the LM3429

ideal for illuminating LEDs in a very diverse, large family of

applications.

Adjustable high-side current sense voltage allows for tight

regulation of the LED current with the highest efficiency pos-

sible. The LM3429 uses Predictive Off-time (PRO) control,

which is a combination of peak current-mode control and a

predictive off-timer. This method of control eases the design

of loop compensation while providing inherent input voltage

feed-forward compensation.

The LM3429 includes a high-voltage startup regulator that

operates over a wide input range of 4.5V to 75V. The internal

PWM controller is designed for adjustable switching frequen-

cies of up to 2.0 MHz, thus enabling compact solutions.

Additional features include analog dimming, PWM dimming,

over-voltage protection, under-voltage lock-out, cycle-by-cy-

cle current limit, and thermal shutdown.

The LM3429 comes in a low profile, thermally efficient TSSOP

EP 14-lead package.

The LM3429Q1 is an Automotive Grade product that is AEC-

Q100 grade 1 qualified

Features

■LM3429Q1 is an Automotive Grade product that is AEC-

Q100 grade 1 qualified (-40°C to +125°C operating

junction temperature)

■VIN range from 4.5V to 75V

■Adjustable current sense voltage

■High-side current sensing

■2Ω, 1A Peak MosFET gate driver

■Input under-voltage protection

■Over-voltage protection

■PWM dimming

■Analog dimming

■Cycle-by-cycle current limit

■Programmable switching frequency

■Thermal Shutdown

Applications

■LED Drivers - Buck, Boost, Buck-Boost, SEPIC

■Indoor and Outdoor SSL

■Automotive

■General Illumination

■Constant-Current Regulators

Typical Boost Application Circuit

300944i9

300944b6

Boost Evaluation Board

9 Series LEDs at 1A

LM3429 LM3429Q1 N-Channel Controller for Constant Current LED Drivers

Connection Diagram

30094404

14-Lead TSSOP EP

Ordering Information

Order Number Spec. Package Type NSC Package

Drawing

Supplied As Features

LM3429MH NOPB TSSOP-14 EP MXA14A 94 Units, Rail

LM3429MHX NOPB TSSOP-14 EP MXA14A 2500 Units, Tape and Reel

LM3429Q1MH NOPB TSSOP-14 EP MXA14A 94 Units, Rail AEC-Q100 Grade 1

qualified. Automotive

Grade Production Flow*

LM3429Q1MHX NOPB TSSOP-14 EP MXA14A 2500 Units, Tape and Reel

*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.

Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified

with the letter Q. For more information go to http://www.national.com/automotive.

Pin Descriptions

Pin Name Description Application Information

1VIN Input Voltage Bypass with 100 nF capacitor to AGND as close to the device as possible in

the circuit board layout.

2 COMP Compensation Connect a capacitor to AGND to set compensation.

3 CSH Current Sense High Connect a resistor to AGND to set signal current. For analog dimming, connect

current source or potentiometer to AGND (see Analog Dimming section).

4 RCT Resistor Capacitor Timing Connect a resistor from the switch node and a capacitor to AGND to set the

switching frequency.

5 AGND Analog Ground Connect to PGND through the DAP copper circuit board pad to provide proper

ground return for CSH, COMP, and RCT.

6 OVP Over-Voltage Protection

Connect to a resistor divider from VOto program output over-voltage lockout

(OVLO). Turn-off threshold is 1.24V and hysteresis for turn-on is provided by

20 µA current source.

7 nDIM Not DIM input

Connect a PWM signal for dimming as detailed in the PWM Dimming section

and/or a resistor divider from VIN to program input under-voltage lockout

(UVLO). Turn-on threshold is 1.24V and hysteresis for turn-off is provided by

20 µA current source.

8 NC No Connection Leave open.

9 PGND Power Ground Connect to AGND through DAP copper pad to provide ground return for GATE.

10 GATE Gate Drive Output Connect to the gate of the external NFET.

11 VCC Internal Regulator Output Bypass with a 2.2 µF–3.3 µF, ceramic capacitor to PGND.

12 IS Main Switch Current Sense Connect to the drain of the main N-channel MosFET switch for RDS-ON sensing

or to a sense resistor installed in the source of the same device.

13 HSP LED Current Sense Positive Connect through a series resistor to LED current sense resistor (positive).

14 HSN LED Current Sense Negative Connect through a series resistor to LED current sense resistor (negative).

DAP

(15) DAP Thermal pad on bottom of IC Connect to AGND and PGND. For thermal considerations see (Note 2).

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LM3429 LM3429Q1

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN, nDIM -0.3V to 76.0V

-1 mA continuous

OVP, HSP, HSN -0.3V to 76.0V

-100 µA continuous

RCT -0.3V to 3.0V

-1 mA to +5 mA continuous

IS -0.3V to 76.0V

-2V for 100 ns

-1 mA continuous

VCC -0.3V to 8.0V

COMP, CSH -0.3V to 6.0V

-200 µA to +200 µA

Continuous

GATE -0.3V to VCC

-2.5V for 100 ns

VCC+2.5V for 100 ns

-1 mA to +1 mA continuous

PGND -0.3V to 0.3V

-2.5V to 2.5V for 100 ns

Maximum Junction

Temperature

Internally Limited

Storage Temperature Range −65°C to +150°C

Maximum Lead Temperature

(Reflow and Solder) (Note 3)

260°C

Continuous Power Dissipation Internally Limited

ESD Susceptibility (Note 4)

Human Body Model 2 kV

Operating Conditions (Note 1)

Operating Junction

Temperature Range −40°C to +125°C

Input Voltage VIN 4.5V to 75V

Electrical Characteristics (Note 1)

Specifications in standard type face are for TJ= 25°C and those with boldface type apply over the full Operating Temperature

Range ( TJ= −40°C to +125°C). Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical

values represent the most likely parametric norm at TJ= +25°C, and are provided for reference purposes only. Unless otherwise

stated the following condition applies: VIN = +14V.

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 6)

Max

(Note 5)Units

STARTUP REGULATOR (VCC)

VCC-REG VCC Regulation ICC = 0 mA 6.30 6.90 7.35 V

ICC-LIM VCC Current Limit VCC = 0V 20 27 mA

IQQuiescent Current Static 1.6 3.0

VCC-UVLO VCC UVLO Threshold VCC Increasing 4.17 4.50

VCC Decreasing 3.70 4.08

VCC-HYS VCC UVLO Hysteresis 0.1

OVER-VOLTAGE PROTECTION (OVP)

VTH-OVP OVP OVLO Threshold OVP Increasing 1.180 1.240 1.280 V

IHYS-OVP OVP Hysteresis Source

Current

OVP Active (high) 10 20 30 µA

ERROR AMPLIFIER

VCSH CSH Reference Voltage With Respect to AGND 1.210 1.235 1.260 V

Error Amplifier Input Bias

Current

-0.6 0 0.6

µA

COMP Sink / Source

Current

10 26 40

Transconductance 100 µA/V

Linear Input Range (Note 7) ±125 mV

Transconductance

Bandwidth

-6dB Unloaded Response

(Note 7)0.5 1.0 MHz

OFF TIMER (RCT)

tOFF-MIN Minimum Off-time RCT = 1V through 1 kΩ35 75 ns

RRCT RCT Reset Pull-down

Resistance

36 120 Ω

VRCT VIN/25 Reference Voltage VIN = 14V 540 565 585 mV

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LM3429 LM3429Q1

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 6)

Max

(Note 5)Units

PWM COMPARATOR

COMP to PWM Offset 700 800 900 mV

CURRENT LIMIT (IS)

VLIM Current Limit Threshold 215 245 275 mV

VLIM Delay to Output 35 75

tON-MIN Leading Edge Blanking

Time

75 250 450

HIGH SIDE TRANSCONDUCTANCE AMPLIFIER

Input Bias Current 10 µA

Transconductance 20 119 mA/V

Input Offset Current -1.5 01.5 µA

Input Offset Voltage -7 07mV

Transconductance

Bandwidth

ICSH = 100 µA

(Note 7)250 500 kHz

GATE DRIVER (GATE)

RSRC(GATE) GATE Sourcing Resistance GATE = High 2.0 6.0 Ω

RSNK(GATE) GATE Sinking Resistance GATE = Low 1.3 4.5

UNDER-VOLTAGE LOCKOUT and DIM INPUT (nDIM)

VTH-nDIM nDIM / UVLO Threshold 1.180 1.240 1.280 V

IHYS-nDIM nDIM Hysteresis Current 10 20 30 µA

THERMAL SHUTDOWN

TSD Thermal Shutdown

Threshold

(Note 7) 165

°C

THYS Thermal Shutdown

Hysteresis

(Note 7) 25

THERMAL RESISTANCE

θJA Junction to Ambient

(Note 2)

14L TSSOP EP 40 °C/W

θJC Junction to Exposed Pad

(DAP)

14L TSSOP EP 5.5 °C/W

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability

and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in

the Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and the device should not be

operated beyond such conditions. All voltages are with respect to the potential at the AGND pin, unless otherwise specified.

Note 2: Junction-to-ambient thermal resistance is highly board-layout dependent. The numbers listed in the table are given for a reference layout wherein the

14L TSSOP EP package has its DAP pad populated with 9 vias. In applications where high maximum power dissipation exists, namely driving a large MosFET

at high switching frequency from a high input voltage, special care must be paid to thermal dissipation issues during board design. In high-power dissipation

applications, the maximum ambient temperature may have to be derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating

junction temperature (TJ-MAX-OP = 125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to ambient thermal resistance

of the package in the application (θJA), as given by the following equation: TA-MAX = TJ-MAX-OP – (θJA × PD-MAX). In most applications there is little need for the full

power dissipation capability of this advanced package. Under these circumstances, no vias would be required and the thermal resistances would be 104 °C/W

for the 14L TSSOP EP. It is possible to conservatively interpolate between the full via count thermal resistance and the no via count thermal resistance with a

straight line to get a thermal resistance for any number of vias in between these two limits.

Note 3: Refer to National’s packaging website for more detailed information and mounting techniques. http://www.national.com/analog/packaging/

Note 4: Human Body Model, applicable std. JESD22-A114-C.

Note 5: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100%

production tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used

to calculate Average Outgoing Quality Level (AOQL).

Note 6: Typical numbers are at 25°C and represent the most likely norm.

Note 7: These electrical parameters are guaranteed by design, and are not verified by test.

Note 8: The measurements were made using the standard buck-boost evaluation board from AN-1985.

Note 9: The measurements were made using the standard boost evaluation board from AN-1986.

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LM3429 LM3429Q1

Typical Performance Characteristics TA=+25°C and VIN = 14V unless otherwise specified

Boost Efficiency vs. Input Voltage

VO= 32V (9 LEDs) (Note 9)

300944b6

Buck-Boost Efficiency vs. Input Voltage

VO= 20V (6 LEDs) (Note 8)

300944b5

Boost LED Current vs. Input Voltage

VO= 32V (9 LEDs) (Note 9)

300944b8

Buck-boost LED Current vs. Input Voltage

VO= 20V (6 LEDs) (Note 8)

300944b7

Analog Dimming

VO= 20V (6 LEDs) (Note 8)

300944b9

PWM Dimming

VO= 20V (6 LEDs) (Note 8)

300944c0

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LM3429 LM3429Q1

VCSH vs. Junction Temperature

300944b0

VCC vs. Junction Temperature

300944b1

VRCT vs. Junction Temperature

300944b2

VLIM vs. Junction Temperature

300944b3

tON-MIN vs. Junction Temperature

300944b4

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LM3429 LM3429Q1

Block Diagram

30094403

Theory of Operation

The LM3429 is an N-channel MosFET (NFET) controller for

buck, boost and buck-boost current regulators which are ideal

for driving LED loads. The controller has wide input voltage

range allowing for regulation of a variety of LED loads. The

high-side differential current sense, with low adjustable

threshold voltage, provides an excellent method for regulating

output current while maintaining high system efficiency. The

LM3429 uses a Predictive Off-time (PRO) control architecture

that allows the regulator to be operated using minimal exter-

nal control loop compensation, while providing an inherent

cycle-by-cycle current limit. The adjustable current sense

threshold provides the capability to amplitude (analog) dim

the LED current and the output enable/disable function allows

for PWM dimming using no external components. When de-

signing, the maximum attainable LED current is not internally

limited because the LM3429 is a controller. Instead it is a

function of the system operating point, component choices,

and switching frequency allowing the LM3429 to easily pro-

vide constant currents up to 5A. This simple controller con-

tains all the features necessary to implement a high efficiency

versatile LED driver.

CURRENT REGULATORS

Current regulators can be designed to accomplish three basic

functions: buck, boost, and buck-boost. All three topologies

in their most basic form contain a main switching MosFET, a

recirculating diode, an inductor and capacitors. The LM3429

is designed to drive a ground referenced NFET which is per-

fect for a standard boost regulator. Buck and buck-boost

regulators, on the other hand, usually have a high-side switch.

When driving an LED load, a ground referenced load is often

not necessary, therefore a ground referenced switch can be

used to drive a floating load instead. The LM3429 can then

be used to drive all three basic topologies as shown in the

Basic Topology Schematics section.

Looking at the buck-boost design, the basic operation of a

current regulator can be analyzed. During the time that the

NFET (Q1) is turned on (tON), the input voltage source stores

energy in the inductor (L1) while the output capacitor (CO)

provides energy to the LED load. When Q1 is turned off

(tOFF), the re-circulating diode (D1) becomes forward biased

and L1 provides energy to both COand the LED load. Figure

1shows the inductor current (iL(t)) waveform for a regulator

operating in CCM.

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LM3429 LM3429Q1

30094498

FIGURE 1. Ideal CCM Regulator Inductor Current iL(t)

The average output LED current (ILED) is proportional to the

average inductor current (IL) , therefore if ILis tightly con-

trolled, ILED will be well regulated. As the system changes

input voltage or output voltage, the ideal duty cycle (D) is var-

ied to regulate ILand ultimately ILED. For any current regulator,

D is a function of the conversion ratio:

Buck

Boost

Buck-boost

PREDICTIVE OFF-TIME (PRO) CONTROL

PRO control is used by the LM3429 to control ILED. It is a

combination of average peak current control and a one-shot

off-timer that varies with input voltage. The LM3429 uses

peak current control to regulate the average LED current

through an array of HBLEDs. This method of control uses a

series resistor in the LED path to sense LED current and can

use either a series resistor in the MosFET path or the MosFET

RDS-ON for both cycle-by-cycle current limit and input voltage

feed forward. D is indirectly controlled by changes in both

tOFF and tON, which vary depending on the operating point.

Even though the off-time control is quasi-hysteretic, the input

voltage proportionality in the off-timer creates an essentially

constant switching frequency over the entire operating range

for boost and buck-boost topologies. The buck topology can

be designed to give constant ripple over either input voltage

or output voltage, however switching frequency is only con-

stant at a specific operating point .

This type of control minimizes the control loop compensation

necessary in many switching regulators, simplifying the de-

sign process. The averaging mechanism in the peak detec-

tion control loop provides extremely accurate LED current

regulation over the entire operating range.

PRO control was designed to mitigate “current mode

instability” (also called “sub-harmonic oscillation”) found in

standard peak current mode control when operating near or

above 50% duty cycles. When using standard peak current

mode control with a fixed switching frequency, this condition

is present, regardless of the topology. However, using a con-

stant off-time approach, current mode instability cannot oc-

cur, enabling easier design and control.

Predictive off-time advantages:

•There is no current mode instability at any duty cycle.

•Higher duty cycles / voltage transformation ratios are

possible, especially in the boost regulator.

The only disadvantage is that synchronization to an external

reference frequency is generally not available.

SWITCHING FREQUENCY

An external resistor (RT) connected between the RCT pin and

the switch node (where D1, Q1, and L1 connect), in combi-

nation with a capacitor (CT) between the RCT and AGND pins,

sets the off-time (tOFF) as shown in Figure 2. For boost and

buck-boost topologies, the VIN proportionality ensures a vir-

tually constant switching frequency (fSW).

30094499

FIGURE 2. Off-timer Circuitry for Boost and Buck-boost

Regulators

For a buck topology, RTand CTare also used to set tOFF,

however the VIN proportionality will not ensure a constant

switching frequency. Instead, constant ripple operation can

be achieved. Changing the connection of RTin Figure 2 from

VSW to VIN will provide a constant ripple over varying VIN.

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LM3429 LM3429Q1

Adding a PNP transistor as shown in Figure 3 will provide

constant ripple over varying VO.

300944a0

FIGURE 3. Off-timer Circuitry for Buck Regulators

The switching frequency is defined:

Buck (Constant Ripple vs. VIN)

Buck (Constant Ripple vs. VO)

Boost and Buck-boost

For all topologies, the CTcapacitor is recommended to be

1 nF and should be located very close to the LM3429.

AVERAGE LED CURRENT

The LM3429 uses an external current sense resistor (RSNS)

placed in series with the LED load to convert the LED current

(ILED) into a voltage (VSNS) as shown in Figure 4. The HSP

and HSN pins are the inputs to the high-side sense amplifier

which are forced to be equal potential (VHSP=VHSN) through

negative feedback. Because of this, the VSNS voltage is forced

across RHSP to generate the signal current (ICSH) which flows

out of the CSH pin and through the RCSH resistor. The error

amplifier will regulate the CSH pin to 1.24V, therefore ICSH can

be calculated:

This means VSNS will be regulated as follows:

ILED can then be calculated:

The selection of the three resistors (RSNS, RCSH, and RHSP) is

not arbitrary. For matching and noise performance, the sug-

gested signal current ICSH is approximately 100 µA. This

current does not flow in the LEDs and will not affect either the

off state LED current or the regulated LED current. ICSH can

be above or below this value, but the high-side amplifier offset

characteristics may be affected slightly. In addition, to mini-

mize the effect of the high-side amplifier voltage offset on LED

current accuracy, the minimum VSNS is suggested to be

50 mV. Finally, a resistor (RHSN = RHSP) should be placed in

series with the HSN pin to cancel out the effects of the input

bias current (~10 µA) of both inputs of the high-side sense

amplifier. Note that he CSH pin can also be used as a low-

side current sense input regulated to the 1.24V. The high-side

sense amplifier is disabled if HSP and HSN are tied to GND.

30094457

FIGURE 4. LED Current Sense Circuitry

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LM3429 LM3429Q1

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